Methods and circuitry for analyzing voltages

ABSTRACT

In circuitry for measuring a voltage at a node, a capacitive divider is coupled to the node, wherein the capacitive divider provides a first output. A resistive divider is coupled to the node, wherein the resistive divider provides a second output.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. patent application Ser. No.15/245,882 filed Aug. 24, 2016, which is fully incorporated herein byreference.

BACKGROUND

Silicon carbide (SiC) based power electronic devices have significantadvantages over silicon (Si) based devices in some respects. Siliconcarbide devices are able to operate at much higher frequencies andtemperatures than silicon devices. In addition, silicon carbide devicesconvert electric power at higher efficiency with lower losses thansilicon devices. Furthermore, silicon carbide devices are capable ofmanaging the same level of power as silicon devices while requiring onlyhalf the size of silicon devices, which enables increases in powerdensity on a circuit board.

DC-to-DC power converters have circuitry or gate controllers that drivegates of transistors, such as metal oxide silicon field-effecttransistors (MOSFETs). The transistors in high voltage and/or high powerconverters pass high current or have to regulate high voltages, whichhas proven to have difficulties.

SUMMARY

Circuitry and methods for measuring the voltage at a node are disclosed.A capacitive divider is coupled to the node, wherein the capacitivedivider provides a first output. A resistive divider is coupled to thenode, wherein the resistive divider provides a second output.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a conventional converter.

FIG. 2A is a graph showing an example of drain voltage of the converterof FIG. 1 as a function of time.

FIG. 2B is a graph showing an example of current flow through thetransistor in the converter of FIG. 1 as a function of time.

FIG. 3 is a schematic diagram of a gate driver coupled to a transistor,wherein the gate driver overcomes the problems with conventional gatedrivers.

FIG. 4 is a detailed block diagram of an example gate driver of FIG. 3.

FIG. 5 is a schematic diagram of an example gate driver of FIG. 4

FIG. 6 is a plurality of timing diagrams showing different voltagefunctions in the circuit of FIG. 5.

FIG. 7 is a flowchart describing the operation of the gate driver ofFIG. 4.

FIG. 8 is a block diagram of an example of the timing extraction anddelay generation circuitry of FIG. 4.

FIG. 9 is a plurality of graphs showing examples of a sampling timingdiagram and a corresponding sensed drain voltage V_(DS) _(_) _(SNS) andgate voltage V_(GS).

FIG. 10 is a plurality of graphs showing the five stages of the drivingtransistor that does not have a Kelvin sensor pin.

FIG. 11 is a plurality of graphs similar to the graphs of FIG. 10, butassociated with a transistor having a Kelvin sensor contact.

FIG. 12 is a flowchart describing an example of a method for convertingthe gate voltage V_(GS) and the drain voltage V_(DS) to digital signals.

FIG. 13 is a block diagram of an example of the modular/adaptive powerstage of FIG. 4.

FIG. 14 is a detailed block diagram of an example modular/adaptive powerstage of FIG. 13.

FIG. 15 is a flowchart describing an example of the operation of thepower stage.

DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS

Silicon carbide (SiC) based power electronic devices have significantadvantages over silicon (Si) based devices in some respects. Siliconcarbide devices operate at much higher frequencies and temperatures thansilicon devices and convert electric power at higher efficiency withlower losses than silicon devices. In addition, silicon carbide devicesare capable of managing the same level of power as silicon devices whilerequiring only half the size of silicon devices, which enables increasesin density on a circuit board. Some conventional devices that requirehigh bandwidth analog drivers are not suitable for fast silicon carbidemetal oxide semiconductor field-effect transistors (MOSFETs) becausethey cannot achieve the required control bandwidth under high currentconditions.

FIG. 1 is a schematic diagram of an exemplary conventional converter100. The converter 100 includes circuitry 104 that drives a switch,which in the example of FIG. 1 is a SiC MOSFET Q10. In other examples,the switch may be a device that has properties similar to the propertiesof a SiC MOSFET or any gate semiconductor switching device. Thecircuitry 104 is fabricated in an integrated circuit having six pins. Aninput 106 receives an input signal, which in the example of FIG. 1 is apulse width modulated (PWM) signal. The converter 100 has an output 108,wherein an output voltage V_(OUT) is present during operation of theconverter 100. The circuitry 104 has a pull-up output 110 and apull-down output 111 that are coupled to a node N10 through tworesistors, R10 and R11. The gate of transistor Q10 is coupled to nodeN10.

The drain of transistor Q10 is coupled to an input voltage V_(IN) andthe source of transistor Q10 is coupled to a negative supply voltage byway of a transistor Q11, which is a similar device to transistor Q10that has a driver (not shown) that is similar to the circuitry 104. Thesource of transistor Q10 is coupled to the converter output 108 by wayof an inductor L10. The converter output 108 is coupled to ground by wayof a capacitor C10. Transistor Q10 conducts current through inductor L10in response to the voltage at node N10. More specifically, the voltageat node N10 determines the drain/source resistance of transistor Q10,which is inversely proportional to the voltage at the gate of transistorQ10. The converter 100 changes the voltage at the gate of transistor Q10causing transistor Q10 to turn off and on, which charges and dischargescapacitor C10.

The values of resistors R10 and R11 are fixed and are selected as atradeoff between efficiency and reducing electromagnetic interference.If the resistance values of resistors R10 and R11 are selected so thattransistor Q10 turns on and off very fast, efficiency is improved.However, the fast transitions in transistor Q10 cause fast changes inthe current flow through transistor Q10, which generates high frequencycomponents at the output 108 and other portions of the converter 100.These high frequency components propagate through the converter 100 andcomponents connected to the output 108 and cause problems with operationof the converter 100 and the connected components.

The converter 100 and similar converters have problems with theovershoot of the drain/source voltage V_(DS) across transistor Q10, therate of change of the drain/source voltage V_(DS), and the rate ofchange of the drain current I_(D). FIG. 2A is a graph 200 showing anexample of the drain/source voltage V_(DS) across transistor Q10 of FIG.1 as a function of time. FIG. 2B is a graph 202 showing an example ofthe drain current I_(D) flowing through transistor Q10 in the converter100 of FIG. 1 as a function of time. The graph 200 shows voltageovershoots 210 and undershoots 212 in the waveform of the drain/sourcevoltage V_(DS). The voltage overshoots 210 are overshoots on the risingedges 214 of the drain/source voltage V_(DS) and the voltage undershoots212 are on the falling edges 216 of the drain/source voltage V_(DS). Thevoltage overshoots 210 and undershoots 212 are the result of inherentinductance and capacitance in the converter 100 coupled with highfrequency shifting of the drain/source voltage V_(DS) and causeinaccuracies in the converter output voltage V_(OUT) in addition to theother above-described problems.

The rate of rise and fall of the rising edge 214 and the falling edge216 is referred to as the slope or dv/dt of the drain/source voltageV_(DS). The rate of rise of the rising edge 214 is referred to as therising slope 220 and the rate of fall of the falling edge 216 isreferred to as the falling slope 222. The slopes are proportional to thehigh frequency components in the waveform of the drain/source voltageV_(DS). The steeper the slopes 220 and 222, the higher the frequencycomponents constituting the waveform. These high frequency componentscan propagate through the converter 100 and devices coupled theretoresulting in electromagnetic interference (EMI). However, the converter100 operates more efficiently with steep slopes 220 and 222, so there istypically a range of slope values where EMI and efficiency are withinpredetermined parameters. Neither EMI nor efficiency are typicallymaximized by the slope values.

The graph 202 of FIG. 2B shows an example of the drain current I_(D)that flows between the drain and the source of transistor Q10 as afunction of time. The waveform of the drain current I_(D) has currentovershoots 230 occurring at the same or the approximate same time aschanges in the drain/source voltage V_(DS) of FIG. 2A. The currentovershoots 230 cause similar problems as the voltage overshoots 210 andundershoots 212 in the waveform of the drain/source voltage V_(DS). Thewaveform of the drain current I_(D) has a falling edge 234 and a risingedge 236. The rate of the change of the drain current I_(D) as afunction of time is referred to as di/dt and is measured as the slope240 of the rising edge 236. As with the slope of the drain/sourcevoltage V_(DS), the slope 240 of the drain current I_(D) should bemaintained within predetermined parameters to maximize efficiency andreduce EMI. As with the drain/source voltage V_(dS), neither EMI norefficiency are maximized when the current I_(D) is maintained within theparameters. The falling edge 234 of the drain current I_(D) may haveovershoots and all the overshoots may be controlled by the methods andcircuitry described herein.

Conventional gate drivers have current sensing components, such as shuntresistors, Desat circuitry, or current mirrors to detect or measure thedrain current I_(D). The current sensing provides information as to thedrain current I_(D), which can indicate a short circuit condition and/oran over current condition. For example, if the converter output 108 iscoupled to ground or a very low resistance, excessive current will flowthrough the converter 100, which could damage the converter 100. Thecurrent sensing components identify the current overshoots 230 and shortcircuit conditions and attempt to rectify the current situation beforedamage occurs. Shunt resistors are current sensing components, but theyare expensive and require large amounts of area. Low resistance shuntresistors typically have an unacceptable inductance to resistance ratio,which makes sensing di/dt difficult. The addition of current mirrors orother devices usually increases the number of pins in an integratedcircuit, which is costly. Desat circuitry requires external circuitryincluding a high voltage diode, which is costly.

The gate drivers and methods of driving gates described herein overcomethe above-described problems associated with shunt resistors and othersensing components. The circuits and method described herein detectand/or measure the voltage on a transistor node (drain or source) underboth high voltage and low voltage conditions. The measurements provideinformation related to the voltage on the node of the transistor fordv/dt and di/dt control and indications as to current flow through thetransistor. The circuits and methods detect over current, such as shortcircuit conditions, without the need of shunt resistors, desatcircuitry, or current mirrors described above.

In summary, the circuits and methods described herein sense current witha resistor divider and a capacitance divider. The resistor dividersenses the on-state voltage across a transistor during an overloadcondition. During the overload condition, the current increases, whichincreases the on-state voltage. Control devices can protect thecircuitry when the on-state voltage is greater than a predeterminedvalue. The capacitive divider senses the drain voltage of the transistoras the transistor is turning on. The drain voltage should drop quicklyas the transistor turns on. If not, there is likely a short within theconverter or some other problem. The converter can be immediately turnedoff without waiting for a blanking time as required in desat circuits.One advantage to the circuits and methods described herein is that thecircuits measure or sense high voltage, but the components within thecircuits do not have to be rated to the high voltages because theyprocess lower voltages that are proportional to the high voltages.

FIG. 3 is a schematic diagram of a gate driver 300 coupled to atransistor Q30, wherein the gate driver 300 overcomes theabove-described problems with conventional gate drivers. The gate driver300 may be part of larger circuitry, such as a voltage converter (notshown). In the examples described herein, transistor Q30 is a siliconcarbide (SiC) MOSFET, but it could be other devices as known in the art,such as an insulated gate bipolar transistor (IGBT). In other examples,transistor Q30 could be any device that has switching performance thatcan be influenced by the voltage and/or current at its control terminal,such as a gate or base of a transistor. The term “gate” and “base” areused interchangeably herein to refer to the gate or base of a transistordepending on the transistor type. The gate driver 300 has a high voltageinput 304 that is coupled to the drain of transistor Q30. The voltage atthe input 304 is the drain/source voltage V_(DS) of transistor Q30,which is also referred to as the drain voltage given that the source oftransistor Q30 is coupled to ground. In some examples, the source oftransistor Q30 is coupled to other devices and not to ground, so thegate controller operates from the potential of the source of transistorQ30 to obtain an accurate drain/source voltage V_(DS). In embodimentswhere transistor Q30 is a P-channel device, the nodes are switched sothat the source is coupled to the high voltage input. The examplesdescribed herein are related to N-channel devices, but they could bealso implemented with P-channel devices by those skilled in the art.

The input 304 is coupled to control and protection circuitry 310, whichis described in detail below. A user input 312 is coupled to thecircuitry 310 and sets operating characteristics of the circuitry 310.The user may be a processor or control module coupled to the gate driver300. A signal input 314 receives a signal for driving the gate oftransistor Q30. In the example of FIG. 3, the signal received at theinput 314 is a pulse width modulated (PWM) signal. A variable driver 320drives the gate of transistor Q30. The driver 320 is coupled to ground;however, in other examples where the transistor Q30 has a Kelvin sensorpin, the driver 320 may be referenced to the Kelvin source pin. Theoutput of the driver 320, which is the gate voltage of transistor Q30,is coupled as an input to the circuitry 310 at a node N31. In someexamples, the driver 320 and the circuitry 310 are fabricated as asingle component.

The circuitry 310 receives user input by way of the input 312. The userinput may be parameters such as high and low values for the dv/dt and/ordi/dt as described above. The circuitry 310 monitors the drain/sourcevoltage V_(DS) of transistor Q30 by way of the input 304 and it alsomonitors the gate voltage V_(GS) of transistor Q30. In some examples,the gate voltage is monitored at node N31 at the output of the driver320 because the voltage at the output of the driver 320 may vary fromthe voltage at the gate of transistor Q30 due to interconnectionimpedance between the devices. The circuitry 310 controls the drive intothe gate of transistor Q30 by way of the driver 320 in response to theabove-described inputs.

FIG. 4 is a block diagram of an example of the gate controller 300 ofFIG. 3. The gate controller 300 includes a high voltage drain sense andcalibration circuit 400 that is referred to herein as the drain sensecircuit 400. The drain sense circuit 400 has an input 402 coupled to thedrain of transistor Q30. Because the source of transistor Q30 is coupledto ground or another node, the voltage at the input 402 is thedrain/source voltage V_(DS) of the transistor Q30. The drain sensecircuit 400 has an output 404 and an output 406. The voltage at theoutput 404 is a sensed V_(DS) voltage V_(DS) _(_) _(SNS) and isproportional to the changes in the drain voltage of transistor Q30 asdescribed in detail below. The output 406 is a voltage V_(ON) _(_)_(SNS), which is proportional to the drain voltage V_(DS) of transistorQ30 during periods when transistor Q30 is on. The drain voltage V_(DS)is reduced to a voltage that may be processed by other components in thegate driver 300 as described below.

In the example of FIG. 4, the output 404 is coupled to a timingextraction and delay generation circuit 410. The circuit 410 processesthe voltage V_(DS) _(_) _(SNS) to determine the timing of the signaldriving the gate of transistor Q30 in addition to other parametersdescribed herein. The circuit 410 is coupled to a dv/dt, di/dt, andovershoot control circuit 416 that analyzes the slope and/or timing ofthe di/dt and/or dv/dt intervals of current and voltage and theovershoots in the transistor Q30. Additionally, the circuit 410 drives amodular/adaptive power stage 420 to drive the gate of transistor Q30 inresponse to the analysis. The power stage 420 drives the gate oftransistor Q30 in a manner to assure that dv/dt, di/dt, and theovershoots are within predetermined specifications. The output 406 iscoupled to a controller 422 that also receives a user input and providesdata to the circuit 416 in response to the voltage V_(ON) _(_) _(SNS)and the user input. In some examples, the output 406 can be fed back tothe circuit 416 to change the operating point of the power stage 420.For example, if during an initial overload current condition, the gatevoltage of transistor Q30 is increased to reduce conduction losses, andif the overload continues to grow, then the gate voltage is reduced toinitiate over current protection turn-off.

FIG. 5 is a schematic illustration of an example of the drain sensecircuit 400 of FIG. 4. The circuit 400 has the input 402 that receivesthe drain or drain/source voltage V_(DS) from transistor Q30 and theoutputs 404 and 406 that output the voltages V_(DS) _(_) _(SNS) andV_(ON) _(_) _(SNS), respectively. The circuit 400 includes a voltagesensing portion 500 and a current sensing portion 502, both of which arecoupled to the input 402. The current sensing portion 502 senses voltagefor other circuitry to determine the current as described below. Thevoltage sensing portion 500 includes a capacitive voltage divider 502,which may be a high ratio capacitive voltage divider. The divider 502includes a capacitor C50, which in some examples is a high voltagecapacitor that is rated to a voltage that is present at the drain oftransistor Q30. The capacitor C50 is coupled in series with a capacitorarray 508 at a node N50. The capacitor array 508 includes a plurality ofcapacitors that are selectively coupled in parallel to node N50. Thecapacitors in the capacitor array 508 are referred to individually ascapacitors C51, C52, C53, and C54. The capacitors are coupled inparallel to node N50 by a plurality of switches. A switch SW52 couplescapacitor C52, a switch SW53 couples capacitor C53, and a switch SW54couples a capacitor C54. The states of the switches SW52-SW54 arecontrolled by calibration circuitry 510. The number of capacitors in thecapacitor array 508 is a design choice and may be greater than or lessthan three capacitors as shown in FIG. 5. In some examples, thecapacitor array 508 has capacitor C50 coupled in series with a singlecapacitor instead of the capacitors and switches in the capacitor array508.

The calibration circuitry 510 has an input 512 coupled to the output ofa comparator 516. A first input of the comparator 516 is coupled to areference voltage V_(REF), which may be set by a user and is set to 5Vin the examples provided herein. A second input of the comparator 516 iscoupled to the output of the capacitor array 508 at node N50 and is thevoltage V_(DS) _(_) _(SNS). The comparator 516 compares the voltageV_(DS) _(_) _(SNS) to the reference voltage V_(REF) and outputs a signalto the input 512 of the calibration circuitry 510 in response to thecomparison. The calibration circuitry 510 iteratively adjusts the statesof the switches SW52-SW54 so that the reference voltage V_(REF) is equalto the voltage V_(DS) _(_) _(SNS) or within a predetermined value of thevoltage V_(DS) _(_) _(SNS). When the switches SW52-SW54 are set so thatthe voltage V_(DS) _(_) _(SNS) is equal to the reference voltageV_(REF), the value of the drain/source voltage V_(DS) is readilydetermined based on the voltage division between the capacitor C50 andthe capacitance of the capacitor array 508.

The current sensing portion 502 includes a resistive divider 530 thatincludes two resistors R50 and R51 coupled at a node N51. The resistivedivider 530 senses the DC current flow through transistor Q30, which maybe used to determine a short circuit or over current situations. Theresistive divider 530 is subject to high voltage with a very steep slopewhen transistor Q30 transitions between off and on states. During theperiods when the input 402 is subject to high voltages, the switch SW58is closed to protect components coupled to the output 406 from the highvoltages. In some examples, resistor R50 and/or resistor R51 have verylow capacitance, such as less than 50 fF. The low capacitance assuresthat resistors R50 and R51 have fast settling times, which is requiredto improve the bandwidth of the drain sense circuit 400. When switchSW58 opens during the on-state of transistor Q30, which is when thedrain of transistor Q30 is a low voltage, the voltage across resistorR51 has to increase from zero to the given divider ratio voltagefraction of the voltage at the input 402. Any parallel/distributedparasitic capacitance in resistor R51 and/or resistor R50 will delay thevoltage increase. To protect coupled circuit components such astransistor Q30, the capacitance is minimized, so as to enable themaximize speed for the voltage rise at the output 406. In some examples,the RC time constant in resistors R50 and R51 is in the range of onemicrosecond or less. One method of improving the time constant is byshielding resistor R50 from ground, such as shielding it with a shieldcoupled to the output 406. Such an embodiment addresses the parasiticcapacitance to the output and improves the settling time.

A sample and hold circuit 532 is coupled to the resistive divider 530and samples and holds the voltage at a node N51 coupled betweenresistors R50 and R51. The circuit 532 includes a holding capacitorC_(HOLD) that is coupled to node N51 by way of switches SW55 and SW56. Abuffer 534 is coupled between the switch SW56 and capacitor C_(HOLD).When the switch SW56 is closed, the circuit 532 samples the voltage atnode N51 onto the capacitor C_(HOLD) by way of the buffer 534. When thevoltage is to be read, the switch SW56 opens and the switch SW55 closesto provide the voltage on capacitor C_(HOLD) at node N51, which is thevoltage V_(ON) _(_) _(SNS).

Clamp control circuitry 540 receives PWM signals for driving the gate oftransistor Q30 and resets the voltage sensing portion 500 and thecurrent sensing portion 502 of the circuitry 400 in response to the PWMsignals. The circuitry 540 has two outputs CLP1 and CLP2 that clamp orshort portions of the voltage sensing portion 500 and the currentsensing portion 502, respectively. The output CLP1 controls switch SW57and the output CLP2 controls switch SW58. Switch SW57 couples node N50to ground when switch SW57 closes and switch SW58 couples node N51 toground when switch SW58 closes. The timing of CLP 1 and CLP2 isdescribed further below.

FIG. 6 is a plurality of timing diagrams showing examples of differentvoltage functions in the circuit 400 of FIG. 5. A graph 600 shows anexample of the drain/source voltage V_(DS), which is the drain/sourcevoltage across transistor Q30, as a function of time. The drain/sourcevoltage V_(DS) rises to a bus voltage V_(BUS), which may be severalthousand volts. The drain/source voltage V_(DS) rises and falls as afunction of the PWM signals. More specifically, when the PWM signalsinstruct transistor Q30 to turn off, the drain/source voltage V_(DS)rises and when the PWM signals instruct transistor Q30 to turn on, thedrain/source voltage V_(DS) decreases. Turning the transistor off or onis sometimes referred to as changing its state. In other examples, thecomplement occurs based on signals decoded by the clamp control 540. Agraph 602 is an exemplary timing diagram of the CLP1 signal as afunction of time and a graph 604 is an exemplary timing diagram of theCLP2 signal as a function of time. As shown in graph 602, the CLP1signal is low during the period that the drain/source voltage V_(DS) ishigh. This low signal opens switch SW57 to generate the voltage V_(DS)_(_) _(SNS) as described below. The CLP2 signal is low during the periodwhen the drain/source voltage V_(DS) is low, which opens switch SW58 andenables the generation of the voltage V_(ON) _(_) _(SNS) as describedbelow.

During periods when the drain/source voltage V_(DS) is high enough suchthat sensing cannot be accomplished on the output 406, the CLP1 signalis low and the CLP2 signal is high, which means the switch SW57 is openand switch SW58 is closed. The drain/source voltage V_(DS) is thendivided between capacitor C50 and the capacitors in the capacitive array508 that are coupled to node N50. The voltage on node N50 is the voltageV_(DS) _(_) _(SNS). Graph 610 is an exemplary timing diagram of thevoltage V_(DS) _(_) _(SNS) over a period of three cycles of thedrain/source voltage V_(DS) as shown in graph 600. In a first iterationor a first cycle shown by graph 612, several capacitors in thecapacitive array 508 are coupled to node N50, so the voltage V_(DS) _(_)_(SNS) is relatively low. More specifically, the voltage V_(DS) _(_)_(SNS) is less than the reference voltage V_(REF). This voltagedifference is measured by the comparator 516, which outputs a signal tothe calibration circuit 510 indicating that the voltage V_(DS) _(_)_(SNS) is less than the reference voltage V_(REF). In response to thesignal from the comparator 516, the calibration circuit 510 opens atleast one switch in the capacitive array 508 to decouple at least onecapacitor in parallel from node N50. During a second iteration when thedrain/source voltage V_(DS) goes high, the reduction in capacitance inthe capacitive array 508 increases the voltage V_(DS) _(_) _(SNS) asshown by graph 614. The comparator 516 generates another signalindicating that the voltage V_(DS) _(_) _(SNS) is still less than thereference voltage V_(REF), so the calibration circuit 510 decouplesanother capacitor from node N50.

During a third iteration of the drain/source voltage V_(DS), thecalibration circuit 510 has coupled less capacitance to node N50. Duringthis third iteration shown by the graph 616, the voltage V_(DS) _(_)_(SNS) is equal to the reference voltage V_(REF) or is within apredetermined amount of the reference voltage V_(REF). The calibrationcircuit 510 or a component associated therewith calculates thedrain/source voltage V_(DS) based on the value of the capacitance in thecapacitive array 508 coupled to node N50. By knowing the value of V_(DS)_(_) _(SNS), which is V_(REF), the value of capacitor C50, and the valueof the capacitance in the capacitive array 508 coupled to node N50, thedrain/source voltage V_(DS) is readily calculated, such as withKirchoff's current law. The drain/source voltage V_(DS) is used by thecircuitry 300, FIG. 4, for many purposes, including feedback to controlthe voltage across transistor Q30. This iterative process may continueduring operation of the circuit 400. In other examples, the sensedvoltage V_(DS) _(_) _(SNS) is measured directly and the number ofcapacitors required to meet the reference voltage is coupled to nodeN50.

The drain/source voltage V_(DS) typically does not settle to zerobetween pulses, but rather stays at some voltage or fluctuates slightly.The current sensing portion 502, FIG. 5, of the circuitry 400 measuresthe drain/source voltage V_(DS) when transistor Q30 is on and producesthe voltage V_(ON) _(_) _(SNS). Transistor Q30 has very littleresistance when it is on, so the drain/source voltage V_(DS) should bevery low, even when it is passing high current. If transistor Q30 isoperating into a short or into a load that pulls excessive current, thedrain/source voltage V_(DS) will be high, or higher than a predeterminedlow voltage value, when transistor Q30 is on. The high drain/sourcevoltage V_(DS) is reflected as a high voltage V_(ON) _(_) _(SNS), whichis processed by the controller 422, FIG. 4, and/or other components. Thecontroller 422 may cause predetermined actions in response to a highdrain/source voltage V_(ON) _(_) _(SNS), such as turning transistor Q30off by way of the modular/adaptive power stage 420. The circuitry 400 isable to detect the overload or short situation in a single pulse of thedrain/source voltage V_(DS). Conventional gate drivers and other devicesfor detecting shorts and overloads take much longer, which may damagedevices and components.

FIG. 7 is a flowchart 700 describing the operation of the gate driver300 of FIG. 4. The method is related to determining the voltage at anode, such as the drain of a transistor. Step 702 of the flowchart 700includes coupling a capacitive divider to the node during a period whena first voltage is at the node. Step 704 includes outputting a firstsignal indicative of the first voltage at the node from the capacitivedivider on a first output. Step 706 includes coupling a resistivedivider to the node during a period when a second voltage is at thenode, wherein the first voltage is greater than the second voltage. Step708 includes outputting a second signal indicative of the second voltageat the node from the resistive divider on a second output.

The gate circuitry 400 of FIG. 4 has been described as being a part ofthe gate driver 300 and operating with other components in the gatedriver 300. In some embodiments, the circuitry 400 operates independentof the gate driver 300 and/or with other gate driver configurations. Forexample, the circuitry 400 may drive the complementary switch in ahalf-bridge converter.

FIG. 8 is a block diagram of an example of the timing extraction anddelay generation circuitry 410 of FIG. 4. The circuitry 410 is beingdescribed herein as operating in conjunction with the other componentsof the gate driver 300 of FIG. 4. However, the circuitry 410 may operateindependent of these components or in conjunction with other gatedrivers. The circuitry 410 serves a plurality of functions includingdetecting the start and end of each turn-on and turn-off stage when thedrain/source voltage V_(DS) rises and falls. This detection enablesevaluation or capturing of the dv/dt and di/dt durations, which are therises and falls of the drain/source voltage V_(DS) and the drain currentI_(D). When the drain/source voltage V_(DS) is analyzed, over-currentevents can also be analyzed and corrected. The circuitry 410 may detectzero voltage switching (ZVS) conditions on transistor Q30, which may bedetected before transistor Q30 turns on. ZVS occurs when the voltage ofthe drain is already low when the gate input is requesting turn-on. Inthis case, the dv/dt and di/dt sensing cannot be achieved and thecontrol of the gate can be changed to accommodate this different mode ofoperation. Typically the gate can then be turned on faster without thelimitations of the adaptive controller.

The circuitry 410 includes a first input 800 that is coupled to the gateof transistor Q30. The voltage at the input 800 is the gate voltageV_(GS) of transistor Q30, which may be the voltage at node N31, FIG. 3.A second input 802 is coupled to the output 404 of FIG. 4 and is thesensed drain voltage V_(DS) _(_) _(SNS), which is proportional to thetransitions in the drain/source voltage V_(DS). In embodiments usingother components, the second input 802 may be coupled to anothercomponent that generates a drain/source voltage or other voltageapplicable for use by the circuitry 410. A third input 806 receives thePWM signal described above. The first input 800 and the second input 802are coupled to inputs of a multiplexrr (MUX) 808. The output of themultiplexor 808 is coupled to a driver 810, which is not included in allexamples of the circuitry 410. The output of the driver 810 is coupledto sample and hold circuitry 814 at a node N81. Some examples of thecircuitry 410 do not include the MUX 808, but rather have two sample andhold circuits, with one coupled to the first input 800 and one coupledto the second input 802.

The sample and hold circuitry 814 includes a plurality of switches 818that selectively couple a plurality of capacitors 820 to node N81. Afirst switch SW81 couples a first capacitor C81 to node N81, a secondswitch SW82 couples a second capacitor C82 to node N81, and an Nthswitch SWN couples a Nth capacitor CN to node N81. The states of theswitches 818 are controlled by a plurality of delays 824 as describedherein. The switches open and close at different times to capture eitherthe voltage V_(DS) _(_) _(SNS) or the gate voltage V_(GS) that ispresent at node N81 depending on which signal is being passed by the MUX808. Node N81 is coupled to the input of an analog-to-digital converter(ADC) 830 that selectively converts the voltages stored on thecapacitors 820 to digital signals.

A waveform analysis and feature extraction unit 834 (sometimes referredto as the analysis circuitry) processes the digital signal generated bythe ADC 830 and controls the drive on transistor Q30 by way of themodular/adaptive power stage 420 as described herein. The functions ofthe unit 834 may be implemented by way of software, hardware, orfirmware. The unit 834 generates signals to control a pulse generator838 and a delay controller 840. The delay controller 840 selectivelyactivates the switches 818 so that one at a time is closed depending onthe sensed drain voltage V_(DS) _(_) _(SNS) or the gate voltage V_(GS).The pulse generator 838 generates sampling pulses to set the samplingperiod that each of the capacitors 820 samples. The pulse generator 838generates a plurality of sampling pulses during the rise and fall periodof the sensed drain voltage V_(DS) _(_) _(SNS) and the gate voltageV_(GS).

The circuitry 410 receives the sensed drain voltage V_(DS) _(_) _(SNS)and the gate voltage V_(GS) at the input of the MUX 808. A signal, suchas a signal generated by the unit 834 instructs the MUX 808 as to whichof the two signals is to be passed by the MUX 808. The signal passed bythe MUX 808 is amplified by the driver 810 and is present at node N81.The switches 818 are initially open and close and open one at a time tosample the voltage at node N81 a total of N times, where N is the numberof capacitors 820 in the sample and hold circuitry 814. In someexamples, the sampling occurs every nanosecond over a period of Nnanoseconds.

FIG. 9 is a plurality of graphs showing examples of the sampling pulses,the sensed drain voltage V_(DS) _(_) _(SNS) and the gate voltage V_(GS).In addition, examples of the sampled sensed voltage V_(DS) _(_) _(SMP)and the sampled gate voltage V_(GS) _(_) _(SMP) are shown in the graphsof FIG. 9. A graph 900 shows an example of the sampling pulses generatedby the pulse generator 838. The pulses must be generated at a frequencyhigh enough to sample any necessary high frequency components of thesensed drain voltage V_(DS) _(_) _(SNS) and the gate voltage V_(GS). Insome embodiments, the sampling pulses have periods of about onenanosecond. By use of the delays 824, sub-nanosecond timing may beachieved without the use of a clock operating in the gigahertz range. Agraph 902 shows an example of the sensed drain voltage V_(DS) _(_)_(SNS) as a dashed line. The sensed drain voltage V_(DS) _(_) _(SNS) issampled by the sample and hold circuitry 814, resulting in the sampleddrain voltage V_(DS) _(_) _(SMP) as show in graph 902. A graph 904 showsan example of the gate voltage V_(GS) as a dashed line and an example ofthe sampled gate voltage V_(GS) _(_) _(SMP). In some examples, thesensed drain voltage V_(DS) _(_) _(SNS) and the gate voltage V_(GS) aresampled at different cycles, so only the single sample and holdcircuitry 814 as shown in FIG. 8 is required. In other examples, onesample and hold circuit samples and holds the sensed drain voltageV_(DS) _(_) _(SNS) and another sample and hold circuit samples and holdsthe gate voltage V_(GS).

The ADC 830 of FIG. 8, measures the voltages on the capacitors 820 shownby the graphs 902 and 904 and converts the analog voltages to digitalsignals, such as binary numbers. The digital signals are input to theunit 834 for processing. Because the sample and hold circuitry 814samples the transitions in the sensed drain voltage V_(DS) _(_) _(SNS),the ADC 830, in some examples, performs the analog-to-digital conversionoutside of the transition periods of the sensed drain voltage V_(DS)_(_) _(SNS).

The unit 834 analyses the digital signals generated by the ADC 830 andgenerates instructions for the power stage 420, FIG. 4, which drivestransistor Q30. The examples described herein control five stages ofdriving transistor Q30 as instructed by the unit 834. Reference is madeto FIG. 10, which is a plurality of graphs showing the five exemplarystages of driving transistor Q30 wherein transistor Q30 does not have aKelvin sensor pin. A first graph 1000 shows example relative drivestrengths on transistor Q30 during the five drive stages, which arereferred to as S1 through S5. A graph 1002 shows an example of thedrain/source voltage V_(DS) in response to the five drive stages S1through S5. A graph 1004 shows the drain current I_(D) during the fivestages. A graph 1006 shows an example of the gate voltage V_(GS) and themaximum gate voltage V_(GS,MAX) during the five stages. The unit 834 orother components may set the periods of the stages S1 through S5.

During the first stage S1, the gate of transistor Q30 is driven hard byinstructions from the unit 834, which turns transistor Q30 on as fast aspossible. During the first stage S1, the drain/source voltage V_(DS) hasnot fallen by an appreciable amount and the drain current I_(D) has notrisen by an appreciable amount. As shown in graph 1006, the gate voltageV_(GS) has started to rise. During subsequent iterations, the strengthof the drive of transistor Q30 may be modified to speed up or slow downthe turn-on rate. During the second stage S2, the slope of the draincurrent I_(D) (di/dt) is controlled. As shown by the graph 1004, thedrain current I_(D) rises fast, but is controlled so as not to exceed apredetermined di/dt limit. The drain/source voltage V_(DS) is maintainedwithin first and second thresholds V_(TH1) and V_(TH2) during the secondstage S2. The third stage S3 focuses on controlling the slope of thedrain voltage dv/dt. As shown by graph 1006, the gate voltage V_(GS)rises slightly in this example and the drain current I_(D) settles. Thedrain/source voltage V_(DS) drops to a third threshold V_(TH3) at theend of the third stage S3. The slope of the drain/source voltage dv/dtis maintained within predetermined limits during the third stage S3. Thefourth stage S4 provides a fast overdrive so as to limit conductionlosses. In this example, the gate voltage V_(GS) has risen during thefourth stage S4. The fifth stage S5 maintains the drain current I_(D),which is noted by holding the gate voltage V_(GS) stable for theremaining time that transistor Q30 is on. The operation of the fifthstage S5 is dependent on feedback from the V_(ON) _(_) _(SNS) voltage atthe output 406 of FIG. 4 and can be adjusted in response to the loadcurrent in a similar manner as described above.

By analyzing the digital data generated by the ADC 830, the unit 834,FIG. 8, is able to identify several parameters related to transistorQ30. In some examples, the unit 834 detects the start and stop of eachof the above-described stages. For example, the di/dt (stage S2) and thedv/dt (stage S3) are determined in addition to the durations of thestages. More specifically, the transition times of di/dt and dv/dt arereadily determined based on the analysis. Over-current events can bedetected by monitoring the drain/source voltage V_(DS) when transistorQ30 is on or turning on. If the drain/source voltage V_(DS) is too high,there is too much current passing through transistor Q30. In a similarmanner, voltage overshoot or undershoot is readily determined. Theanalysis further enables the unit 834 to adjust the timing and power ofthe drive signal applied to the gate of transistor Q30 during all thestages described above. A zero voltage switching (ZVS) condition is alsodetermined before transistor Q30 is turned on in some applications. AZVS condition occurs when the drain voltage is low and the gate input isrequesting that the transistor turn on. In ZVS conditions, dv/dt anddi/dt sensing cannot be achieved and the control of the gate can bechanged to accommodate this different mode of operation. Typically, thetransistor can be turned on faster without the limitations of anadaptive controller.

FIG. 11 is a plurality of graphs similar to the graphs of FIG. 10, butassociated with a transistor having a Kelvin source contact. The Kelvinsource contact provides a low impedance contact to the source contact oftransistor Q30. The parameters related to transistor Q30 differ slightlywhen transistor Q30 has a Kelvin source contact. A graph 1100 shows thedrain/source voltage V_(DS) of transistor Q30. As shown, the voltagedifference between the first voltage threshold V_(TH1) and the secondvoltage threshold V_(TH2) is greater than examples where transistor Q30does not include a Kelvin source contact. A graph 1102 shows the draincurrent I_(D) flowing through transistor Q30. A portion 1104 of thegraph 1102 shows a steeper di/dt, which is indicative of the Kelvinsource. A graph 1106 shows the gate voltage V_(DS) of transistor Q30. Aportion 1108 shows a reduced kickback from the source inductance intransistor Q30. By analyzing the drain/source voltage V_(DS), the draincurrent I_(D), and the gate voltage V_(DS), the unit 834 or othercomponents can determine whether transistor Q30 has a Kelvin sourcecontact.

In some examples, the pulse generator 838 can adjust the pulses to belonger or shorter to increase the resolution of the time base of thesampling in accordance with the speed of the device being switched. Forexample, a slow IGBT may need a longer time between samples than a SiCdevice, but will only have a fixed number of sample points to sample theshow transition waveform.

FIG. 12 is a flowchart 1200 describing an example of a method forconverting the gate voltage V_(GS) and the drain/source voltage V_(DS)to digital signals. It is noted that the circuits and methods may beapplied to converting voltages other than drain/source and gate voltagesto digital signals. Step 1202 of the flowchart 1200 includes coupling aplurality of capacitors to a node one at a time, wherein the signalbeing sampled is present at the node. Step 1204 includes measuring thevoltage of a first capacitor in the plurality of capacitors. Step 1206includes converting the measured voltage of the first capacitor to adigital signal. Step 1208 includes measuring the voltage of a secondcapacitor in the plurality of capacitors. Step 1210 includes convertingthe measured voltage of the second capacitor to a digital signal.

FIG. 13 is a block diagram of an example of the modular/adaptive powerstage 420 of FIG. 4. The power stage 420 may be used in gate driversthat differ from the gate driver 300 described in FIGS. 3 and 4. Thepower stage 420 includes a driver stage 1300 having a plurality ofindividual driver slices. In the example of FIG. 13, there are twodriver slices, a first driver slice 1302 and a second driver slice 1304.The driver slices 1302 and 1304 sum current to drive the gate oftransistor Q30 as described below. The driver slices 1302 and 1304 maybe binary weighted, thermometer weighted, or weighted based on otherfactors. The conductors are illustrated in FIG. 13 as being singleconductors. However, the conductors may be a plurality of parallelconductors, such as bus lines.

The first driver slice 1302 is similar to the second driver slice 1304and other driver slices that may be included in the power stage 420. Thefollowing description of the first driver slice 1302 is applicable toall the aforementioned driver slices. The first driver slice 1302includes a first transistor Q131 and a second transistor Q132 that arecoupled in series at a node N131. The transistors Q131 and Q132 may beNMOS, PMOS, BJT or other transistors known to those skilled in the art.Node 131 is coupled to the gate of transistor Q30, so the voltage atnode N131 is the gate voltage V_(GS) of transistor Q30. Transistors Q131and Q132 are coupled between a power supply 1310 and a ground node,wherein the power supply 1310 supplies a voltage V131. The ground nodeis at a potential that is different than the voltage V131 supplied bythe power supply 1310. In some examples, the voltage V131 and the groundnode may be varying supplies.

A power supply 1312 is coupled to the power supply 1310 and generates avoltage V132 in examples where the transistor Q131 is an NMOS pull-upimplementation. In the example of FIG. 13, the power supply 1312 is abootstrap supply. The output of the supply 1312 is coupled across acapacitor C131 and serves to supply power to a driver 1316 that drivesthe gate of transistor Q131. The power supply 1312 enables the gate oftransistor Q131 to be driven at a different voltage than the voltageV131 supplied to the drain of transistor Q131. The input of the driver1316 is coupled to a shift register 1320. The shift register 1320 storesthe drive power in the different drive stages (S1-S5) for transistor Q30as described above. The shift register 1320 outputs signals to turn onspecific numbers of driver slices during different stages of the driveof transistor Q30. For example, if transistor Q30 is to be driven strongduring one stage, several drive slices will be active during that stage.Likewise, if transistor Q30 is to be driven light during another stage,a few drive slices will be active during that stage.

FIG. 14 is a partial detailed block diagram of an example implementationof the modular/adaptive power stage 420 of FIG. 13. The shift register1320 in the example power stage 420 of FIG. 14 includes a plurality ofD/Q flip-flops 1400, referenced individually as S1, S2, S3, S4, and S5,coupled in series. The individual flip-flops S1-S5 correlate to the fivedrive stages S1-S5 described above and may vary depending on the numberof stages. The flip-flops 1400 have four channels or bits to pass fromthe input to the output, which enables a selection of sixteen discretepower levels driving transistor Q30 during each of the drive stagesS1-S5. In other examples, the shift register 1320 has different numbersof channels to enable different numbers of discrete power levels drivingtransistor Q30. Each of the flip-flops 1400 shown in FIG. 14 mayactually have one flip-flop per channel. For example, the flip-flop S1may actually be four flip-flops, one per channel, functioning in unison.

The flip-flops 1400 are preset with the drive power corresponding totheir respective stage in the drive cycle by a preset controller 1404.For example, if the first drive stage S1 is to drive transistor Q30strong, a large number is preloaded into flip-flop S1. Numberscorresponding to the remaining drive stages S2-S5 are preloaded intotheir respective flip-flops. The flip-flops 1400 are driven by a PWMgenerator 1408. The pulse widths generated by the PWM generatorcorrelate to the time each of the drive stages into transistor Q30 is tobe active as described below.

The driver stage 1300 of FIG. 14 includes a plurality of driver slicesas described with reference to FIG. 13. The drive stage 1300 includesfour drive slices 1410, one for each channel. The drive slices 1410 arereferenced individually as the first drive slice 1411, the second driveslice 1412, the third drive slice 1413, and the fourth drive slice 1414.The drive slices 1410 drive the gate of transistor Q30 at differentdrive levels. In the example of FIG. 14, the second drive slice 1412drives twice as hard as the first drive slice 1411. The third driveslice 1413 drives twice as hard as the second drive slice 1412, and thefourth drive slice 1414 drives twice as hard as the third drive slice1413. For example, the first driver slice 1411 may have a singletransistor pair, the second drive slice 1412 may have two transistorpairs, the third drive slice 1413 may have four transistor pairs, andthe fourth drive slice 1414 may have eight transistor pairs. Other driveconfigurations may be used and is determined by design choice.

The power stage 420 operates by receiving data regarding the drive powerto transistor Q30 for each drive stage. In the example of FIG. 4, thesedrive stages are set by the timing extraction and delay generation 410,FIG. 4, and/or the dv/dt, di/dt, and overshoot control 416. These drivevalues are loaded into their corresponding flip-flops 1400. The powerstage 420 also receives information as the amount of time each drivestage S1-S5 drives transistor Q30. During the first drive stage S1, thedrive value of the first stage that was loaded into the first flip-flopS1 is output to the drive stage 1300. The drive value is a digitalnumber wherein the least significant bit is coupled to the first driveslice 1411 and the most significant bit is coupled to the fourth driveslice 1414. This configuration enables larger drive values to turn onmore transistors in the drive stage 1300, which drives transistor Q30harder. The drive value into the drive stage 1300 remains for the periodset by the PWM generator 1408 and as dictated by the drive time of thefirst drive stage. The timing is generated in the controller 422, FIG.4, and/or the control 416 based on information extracted in thecircuitry 410.

When the first drive stage has output to the drive stage 1300, the shiftregister 1320 shifts all the values stored in the shift register 1320.The drive value of the second drive stage is then stored in the firstflip-flop S1. When the period for the first drive value has expired, theshift register 1320 shifts the second drive value to the drive stage1300. The drive slices 1410 are activated per the second drive value asdescribed above with reference to the first drive value. The processcontinues through all the drive stages S1-S5. When the last drive valueis output to the drive slices 1410, another set of driver values isloaded into the shift register 1320 and the process continues. Thevalues loaded into the shift register 1320 will typically change toconverge the parameters of transistor Q30 to specific values.

The driver stage 1300 is controlled adaptively to turn on a specificnumber of driver slices 1410 at a given time, which enables variabilityin the drive strength and time. In some examples, the drive slices 1410can function as resistors or as constant current sources. Thus, thedrive strength can be either a varying current or a varying resistance(impedance).

In some embodiments, the voltage V131, FIG. 13, may be a controlledvariable. Thus the voltage to switch the gates can be controlled inamplitude as well as time. Such embodiments enable a higher gate voltageduring the di/dt portion of turn-on to improve switching speed in thepresence of common source inductance and/or reduce gate voltage duringturn-off in overload to reduce the drain current and minimize voltageovershoot. Generating the variable gate voltage may be achieved by usinga higher external supply voltage and the variable supply is internallygenerated through a linear regulator or similar device. The trade-offwith a fixed voltage driver is increased power loss in the driver inexchange for improved losses in the transistors in drive slices 1410 aswell as improved control of the drive stage 1300 during faultconditions.

FIG. 15 is a flowchart 1500 describing an example of the operation ofthe power stage 420. More specifically, the flowchart 1550 describes amethod for driving a device at different power levels during differentdrive stages of a drive cycle. Step 1502 of the flowchart 1500 includesreceiving power level information related to the power levels for theplurality of drive stages. Step 1504 includes receiving timinginformation related to the length of each drive stage. Step 1506includes driving the device at a first power level during a first drivestage lasting a first period. Step 1508 includes driving the device at asecond power level during a second drive stage lasting a second period.

Modifications are possible in the described embodiments, and otherembodiments are possible, within the scope of the claims.

What is claimed is:
 1. Circuitry for analyzing a voltage at a node, thecircuitry comprising: a capacitor divider coupled to the node, thecapacitor divider having a first output, wherein voltages at the firstoutput are proportional to first voltages at the node; a resistivedivider coupled to the node, the resistive divider having a secondoutput, wherein voltages at the second output are proportional to secondvoltages at the node, the second voltages being different from the firstvoltages; first switching circuitry coupled to the capacitor dividerfor: activating the capacitive divider when the first voltages are atthe node; and disabling the capacitor divider when the first voltagesare not at the node; and second switching circuitry coupled to theresistive divider for: activating the resistive divider when the secondvoltages are at the node; and disabling the resistive divider when thesecond voltages are not at the node.
 2. The circuitry of claim 1,wherein the capacitor divider includes a first capacitor coupled betweenthe node and the first output and at least one second capacitor coupledto the first output.
 3. The circuitry of claim 1, wherein the capacitordivider includes a first capacitor coupled between the node and thefirst output and a plurality of second capacitors coupled to the firstoutput by switches for selectively coupling individual capacitors in theplurality of second capacitors to the first output.
 4. The circuitry ofclaim 3, wherein the plurality of second capacitors are selectivelycoupled to the first output to obtain a predetermined voltage at thefirst output.
 5. The circuitry of claim 4, further comprising circuitryfor determining the voltage at the node in response to the capacitanceselectively coupled to the first output.
 6. The circuitry of claim 5,further comprising a comparator for comparing the voltage at the firstoutput to a predetermined voltage, the output of the comparator beingcoupled to the circuitry for determining the voltage at the firstoutput.
 7. The circuitry of claim 1, further comprising a sample andhold circuit coupled to the second output.
 8. The circuitry of claim 1further comprising a clamp control coupled to the first switchingcircuitry and the second switching circuitry for controlling the statesof the first switching circuitry and the second switching circuitry inresponse to an input signal.
 9. The circuitry of claim 8, wherein theinput signal is a pulse width modulated signal.
 10. The circuitry ofclaim 1, wherein the node is coupled to one of a source or drain of atransistor.
 11. The circuitry of claim 10, wherein the gate of thetransistor is driven at least partially in response to the voltage atthe first output and the voltage at the second output.
 12. A method ofdetermining a voltage at a node, the method comprising: coupling acapacitive divider to the node during a period when a first voltage isat the node and decoupling the capacitive divider from the node when thefirst voltage is not at the node; outputting a first signal indicativeof the first voltage at the node from the capacitive divider on a firstoutput; coupling a resistive divider to the node during a period when asecond voltage is at the node, wherein the first voltage is greater thanthe second voltage, and decoupling the resistive divider from the nodewhen the second voltage is not at the node; outputting a second signalindicative of the second voltage at the node from the resistive divideron a second output.
 13. The method of claim 12 wherein outputtingsignals indicative of the first and second voltages at the node includesoutputting signals from successive first voltages and second voltages onthe node.
 14. The method of claim 12, wherein outputting a first signalindicative of the first voltage at the node from the capacitive divideron a first output comprises: coupling a first capacitor between the nodeand the first output; coupling at least one second capacitor to thefirst output; measuring the voltage at the first output; changing thecapacitance of the at least one second capacitor until the voltage atthe first output is a predetermined voltage; and determining the voltageat the node in response to the value of the first capacitor and thevalue of the at least one second capacitor.
 15. The method of claim 12,wherein outputting a first signal indicative of the first voltage at thenode from the capacitive divider on a first output comprises: coupling afirst capacitor between the node and the first output; coupling at leastone second capacitor to the first output; measuring the voltage at thefirst output; changing the capacitance of the at least one secondcapacitor continually during subsequent periods when the first voltageis present at the node until the voltage at the first output is apredetermined voltage; and determining the voltage at the node inresponse to the value of the first capacitor and the value of the atleast one second capacitor.
 16. The method of claim 12, wherein the nodeis coupled to one of either the drain or source of a transistor.
 17. Themethod of claim 16, further comprising changing the drive on the gate ofthe transistor in response to at least one of the first signal or thesecond signal.
 18. The method of claim 12, wherein coupling thecapacitive divider to the node and coupling the resistive divider to thenode comprises coupling one of either the resistive divider to the nodeor coupling the capacitive divider to the node.
 19. Circuitry formeasuring a voltage across a transistor during voltage transitionperiods and periods of DC voltage across the transistor, the circuitrycomprising: a node coupled to one of either the source or drain of thetransistor; a capacitor divider coupled to the node, the capacitordivider having a first output, wherein voltages at the first output areproportional to voltage changes on the node, the capacitive dividerhaving a first capacitor coupled between the node and the first outputand at least one second capacitor selectively coupled to the firstoutput; selection circuitry for selectively coupling the at least onesecond capacitor to the first output in response to the voltage of thefirst output; a resistive divider coupled to the node, the resistivedivider coupled to a second output, wherein voltages at the secondoutput are proportional to DC voltages on the node; first switchingcircuitry coupled to the capacitor divider for: activating thecapacitive divider when first voltages are at the node; and disablingthe capacitive divider when the first voltages not at the node; andsecond switching circuitry coupled to the resistive divider for:activating the resistive divider when second voltages different from thefirst voltages are at the node; and disabling the resistive divider whenthe second voltages are not at the node; wherein one of the firstswitching circuitry or the second switching circuitry activates one ofthe capacitor divider or the resistive divider at a time.
 20. Thecircuitry of claim 19, wherein the selection circuitry couples the atleast one second capacitor to the first output to achieve apredetermined output at the first output.